ANALOG DEVICES AD8013

Single Supply, Low Power, a Triple Video Amplifier AD8013 FEATURES PIN CONFIGURATION Three Video Amplifiers in One Package 14-Pin DIP & SOIC Package Drives Large Capacitive Load Excellent Video Specifications (R = 150 V) L DISABLE 1 1 14 OUT 2 Gain Flatness 0.1 dB to 60 MHz 0.02% Differential Gain Error DISABLE 2 2 13 –IN 2 0.06° Differential Phase Error DISABLE 3 3 12 +IN 2 Low Power 4 +V 11 –V AD8013 S S Operates on Single +5 V to +13 V Power Supplies +IN 1 5 10 +IN 3 4 mA/Amplifier Max Power Supply Current –IN 1 6 9 –IN 3 High Speed OUT 1 7 8 OUT 3 140 MHz Unity Gain Bandwidth (3 dB) Fast Settling Time of 18 ns (0.1%) 1000 V/ms Slew Rate High Speed Disable Function per Channel differential gain and phase error of 0.02% and 0.06°. This Turn-Off Time 30 ns makes the AD8013 ideal for broadcast and professional video Easy to Use electronics. 95 mA Short Circuit Current Output Swing to Within 1 V of Rails The AD8013 offers low power of 4 mA per amplifier max and runs on a single +5 V to +13 V power supply. The outputs of APPLICATIONS each amplifier swing to within one volt of either supply rail to LCD Displays easily accommodate video signals. The AD8013 is unique Video Line Driver among current feedback op amps by virtue of its large capacitive Broadcast and Professional Video load drive. Each op amp is capable of driving large capacitive Computer Video Plug-In Boards loads while still achieving rapid settling time. For instance it Consumer Video can settle in 18 ns driving a resistive load, and achieves 40 ns RGB Amplifier in Component Systems (0.1%) settling while driving 200 pF. PRODUCT DESCRIPTION The outstanding bandwidth of 140 MHz along with 1000 V/μs The AD8013 is a low power, single supply, triple video of slew rate make the AD8013 useful in many general purpose amplifier. Each of the three amplifiers has 30 mA of output high speed applications where a single +5 V or dual power current, and is optimized for driving one back terminated video supplies up to ±6.5 V are required. Furthermore the AD8013’s load (150 Ω) each. Each amplifier is a current feedback amp- high speed disable function can be used to power down the lifier and features gain flatness of 0.1 dB to 60 MHz while offering amplifier or to put the output in a high impedance state. This can then be used in video multiplexing applications. The AD8013 is available in the industrial temperature range of G = +2 0.2 –40°C to +85°C. R = 150Ω L 0.1 500mV 0 500ns V = ± 5V S 100 –0.1 9 0 V = +5V S –0.2 –0.3 –0.4 1 0 0% –0.5 5V 1G 1M 10M 100M FREQUENCY – Hz Channel Switching Characteristics for a 3:1 Mux Fine-Scale Gain Flatness vs. Frequency, G = +2, R = 150 Ω L REV. A Information furnished by Analog Devices is believed to be accurate and © Analog Devices, Inc., 1995 reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Tel: 617/329-4700 Fax: 617/326-8703 NORMALIZED GAIN – dB (@ T = +258C, R = 150 V, unless otherwise noted) AD8013–SPECIFICATIONS A LOAD Model AD8013A Conditions V Min Typ Max Units S DYNAMIC PERFORMANCE Bandwidth (3 dB) No Peaking, G = +2 +5 V 100 125 MHz No Peaking, G = +2 ±5 V 110 140 MHz Bandwidth (0.1 dB) No Peaking, G = +2 +5 V 50 MHz No Peaking, G = +2 ±5 V 60 MHz Slew Rate 2 V Step +5 V 400 V/μs 6 V Step ±5 V 600 1000 V/μs Settling Time to 0.1% 0 V to +2 V ±5 V 18 ns 4.5 V Step, C = 200 pF ±6 V 40 ns LOAD R > 1 kΩ, R = 4 kΩ LOAD FB NOISE/HARMONIC PERFORMANCE Total Harmonic Distortion f = 5 MHz, R = 1 k ±5 V –76 dBc C L f = 5 MHz, R = 150 Ω±5 V –66 dBc C L Input Voltage Noise f = 10 kHz +5 V, ±5 V 3.5 nV/√Hz Input Current Noise f = 10 kHz (–I ) +5 V, ±5 V 12 pA/√Hz IN 1 Differential Gain (R = 150 Ω) f = 3.58 MHz, G = +2 +5 V 0.05 % L ±5 V 0.02 0.05 % 1 Differential Phase (R = 150 Ω) f = 3.58 MHz, G = +2 +5 V 0.06 Degrees L ±5 V 0.06 0.12 Degrees DC PERFORMANCE Input Offset Voltage T to T +5 V, ±5 V 2 5 mV MIN MAX Offset Drift 7 μV/°C Input Bias Current (–) +5 V, ±5 V 2 10 μA Input Bias Current (+) T to T +5 V, ±5 V 3 15 μA MIN MAX Open-Loop Transresistance +5 V 650 800 kΩ T to T 550 kΩ MIN MAX ±5 V 800 k 1.1 M Ω T to T 650 kΩ MIN MAX INPUT CHARACTERISTICS Input Resistance +Input ±5 V 200 kΩ –Input ±5 V 150 Ω Input Capacitance ±5 V 2 pF Input Common-Mode Voltage Range ±5 V 3.8 ±V +5 V 1.2 3.8 +V Common-Mode Rejection Ratio Input Offset Voltage +5 V 52 56 dB Input Offset Voltage ±5 V 52 56 dB –Input Current +5 V, ±5 V 0.2 0.4 μA/V +Input Current +5 V, ±5 V 5 7 μA/V OUTPUT CHARACTERISTICS Output Voltage Swing R = 1 kΩ V –V 0.8 1.0 V L OL EE V –V 0.8 1.0 V CC OH R = 150 Ω V –V 1.1 1.3 V L OL EE V –V 1.1 1.3 V CC OH Output Current +5 V 30 mA ±5 V 25 30 mA Short-Circuit Current ±5 V 95 mA Capacitive Load Drive ±5 V 1000 pF MATCHING CHARACTERISTICS Dynamic Crosstalk G = +2, f = 5 MHz +5 V, ±5 V 70 dB Gain Flatness Match f = 20 MHz ±5 V 0.1 dB DC Input Offset Voltage +5 V, ±5 V 0.3 mV –Input Bias Current +5 V, ±5 V 1.0 μA –2– REV. A AD8013 Model AD8013A Conditions V Min Typ Max Units S POWER SUPPLY Operating Range Single Supply +4.2 +13 V Dual Supply ±2.1 ±6.5 V Quiescent Current/Amplifier +5 V 3.0 3.5 mA ±5 V 3.4 4.0 mA ±6.5 V 3.5 mA Quiescent Current/Amplifier Power Down +5 V 0.25 0.35 mA ±5 V 0.3 0.4 mA Power Supply Rejection Ratio Input Offset Voltage V = ±2.5 V to ±5 V 70 76 dB S –Input Current +5 V, ±5 V 0.03 0.2 μA/V +Input Current +5 V, ±5 V 0.07 1.0 μA/V DISABLE CHARACTERISTICS Off Isolation f = 6 MHz +5 V, ±5 V –70 dB Off Output Impedance G = +1 +5 V, ±5 V 12 pF Turn-On Time 50 ns Turn-Off Time 30 ns Switching Threshold –V + xV 1.3 1.6 1.9 V S NOTES 1 The test circuit for differential gain and phase measurements on a +5 V supply is ac coupled. Specifications subject to change without notice. 1 Maximum Power Dissipation ABSOLUTE MAXIMUM RATINGS The maximum power that can be safely dissipated by the AD8013 Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . 13.2 V Total 2 is limited by the associated rise in junction temperature. The Internal Power Dissipation maximum safe junction temperature for the plastic encapsulated Plastic (N) . . . . . . . . . 1.6 Watts (Observe Derating Curves) parts is determined by the glass transition temperature of the Small Outline (R) . . . . 1.0 Watts (Observe Derating Curves) plastic, about 150°C. Exceeding this limit temporarily may Input Voltage (Common Mode) . . Lower of ±V or ±12.25 V S cause a shift in parametric performance due to a change in the Differential Input Voltage . . . . . . . . Output ±6 V (Clamped) stresses exerted on the die by the package. Exceeding a junction Output Voltage Limit temperature of 175°C for an extended period can result in Maximum . . . . . . . . . Lower of (+12 V from –V ) or (+V ) S S device failure. Minimum . . . . . . . . . Higher of (–12.5 V from +V ) or (–V ) S S Output Short Circuit Duration While the AD8013 is internally short circuit protected, this may . . . . . . . . . . . . . . . . . . . . Observe Power Derating Curves not be enough to guarantee that the maximum junction temper- Storage Temperature Range ature is not exceeded under all conditions. To ensure proper N and R Package . . . . . . . . . . . . . . . . . . . –65°C to +125°C operation, it is important to observe the derating curves. Operating Temperature Range It must also be noted that in (noninverting) gain configurations AD8013A . . . . . . . . . . . . . . . . . . . . . . . . . . –40°C to +85°C (with low values of gain resistor), a high level of input overdrive Lead Temperature Range (Soldering 10 sec) . . . . . . . . +300°C can result in a large input error current, which may result in a NOTES significant power dissipation in the input stage. This power 1 Stresses above those listed under “Absolute Maximum Ratings” may cause must be included when computing the junction temperature rise permanent damage to the device. This is a stress rating only and functional operation of the device at these or any other conditions above those indicated in due to total internal power. the operational section of this specification is not implied. Exposure to absolute 2.5 maximum rating conditions for extended periods may affect device reliability. 2 T = +150°C Specification is for device in free air: J 14-Pin Plastic DIP Package: θ = 75°C/Watt JA 14-Pin SOIC Package: θ = 120°C/Watt JA 2.0 14-PIN DIP PACKAGE ORDERING GUIDE 1.5 Temperature Package Package Model Range Description Options 14-PIN SOIC AD8013AN –40°C to +85°C 14-Pin Plastic DIP N-14 1.0 AD8013AR-14 –40°C to +85°C 14-Pin Plastic SOIC R-14 AD8013AR-14-REEL –40°C to +85°C 14-Pin Plastic SOIC R-14 AD8013AR-14-REEL7 –40°C to +85°C 14-Pin Plastic SOIC R-14 0.5 AD8013ACHIPS –40°C to +85°C Die Form –50 –40 –30 –20 –10 0 10 20 30 40 50 60 70 80 90 AMBIENT TEMPERATURE – °C Maximum Power Dissipation vs. Ambient Temperature –3– REV. A MAXIMUM POWER DISSIPATION – Watts AD8013 METALIZATION PHOTO Contact factory for latest dimensions. Dimensions shown in inches and (mm). +v +IN1 s DISABLE 3 5 3 4 –IN1 6 2 DISABLE 2 1 DISABLE 1 OUT1 7 0.044 (1.13) 14 OUT 2 OUT3 8 –IN3 9 10 12 13 11 +IN3 +IN2 –IN2 –V S 0.071 (1.81) CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily WARNING! accumulate on the human body and test equipment and can discharge without detection. Although the AD8013 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recom- ESD SENSITIVE DEVICE mended to avoid performance degradation or loss of functionality. 6 12 5 10 NO LOAD 4 8 R = 150Ω L 3 6 2 4 1 2 0 0 172 3456 172 3456 SUPPLY VOLTAGE – ± Volts SUPPLY VOLTAGE – ± Volts Figure 1. Input Common-Mode Voltage Range vs. Figure 2. Output Voltage Swing vs. Supply Voltage Supply Voltage –4– REV. A COMMON-MODE VOLTAGE RANGE – ± Volts OUTPUT VOLTAGE SWING – V p-p AD8013 10 3 V = ±5V S 2 8 1 6 0 –I B 4 –1 V = +5V S 2 –2 +I B 0 –3 10 100 1k 10k –60 –40 –20 0 20 40 60 80 100 120 140 LOAD RESISTANCE – Ω JUNCTION TEMPERATURE – °C Figure 3. Output Voltage Swing vs. Load Resistance Figure 6. Input Bias Current vs. Junction Temperature 2 12 1 11 V = ± 5V S 0 10 –1 9 V = +5V S V = +5V S –2 8 V = ±5V S –3 7 –4 6 –60 –40 –20 0 20 40 60 80 100 120 140 –60 –40 –20 0 20 40 60 80 100 120 140 JUNCTION TEMPERATURE – °C JUNCTION TEMPERATURE – °C Figure 4. Total Supply Current vs. Junction Temperature Figure 7. Input Offset Voltage vs. Junction Temperature 140 11 V = ± 5V S T = +25°C 130 A 10 SOURCE 120 9 SINK 100 8 90 80 7 –60 –40 –20 0 20 40 60 80 100 120 140 17 23456 JUNCTION TEMPERATURE – °C SUPPLY VOLTAGE – ± Volts Figure 5. Supply Current vs. Supply Voltage Figure 8. Short Circuit Current vs. Junction Temperature REV. A –5– SUPPLY CURRENT – mA OUTPUT VOLTAGE SWING – V p-p SUPPLY CURRENT – mA SHORT CIRCUIT CURRENT – mA INPUT OFFSET VOLTAGE – mV INPUT BIAS CURRENT – µA AD8013 1k 70 G = +2 R R V 60 CM 100 R R 50 10 40 V = ±5V S 1 30 0.1 20 10 0.01 100k 1M 10M 100M 1G 100k 1M 10M 100M 1G FREQUENCY – Hz FREQUENCY – Hz Figure 9. Closed-Loop Output Resistance vs. Figure 12. Common-Mode Rejection vs. Frequency Frequency 100k 80 V = ±5V S 70 10k 60 50 +PSR 1k 40 30 –PSR 100 20 10 0 10 1M 10M 100M 1G 100k 1M 10M 100M 1G FREQUENCY – Hz FREQUENCY – Hz Figure 10. Output Resistance vs. Frequency, Disabled Figure 13. Power Supply Rejection Ratio vs. Frequency State 1k 1k 140 0 V = ±5V S –45 R = 1k L –90 120 –135 100 100 100 –180 NONINVERTING I 80 10 10 INVERTING I 60 V NOISE 1 1 40 100 1k 10k 100k 1M 10k 100k 1M 10M 100M 1G FREQUENCY – Hz FREQUENCY – Hz Figure 11. Input Current and Voltage Noise vs. Figure 14. Open-Loop Transimpedance vs. Frequency Frequency (Relative to 1 Ω) –6– REV. A VOLTAGE NOISE nV/√Hz OUTPUT RESISTANCE – Ω CLOSED-LOOP OUTPUT RESISTANCE – Ω CURRENT NOISE pA/√Hz TRANSIMPEDANCE – dB POWER SUPPLY REJECTION – dB COMMON-MODE REJECTION – dB PHASE – Degrees AD8013 –30 G = +1 G = +2 R = 150Ω –40 0 V = 2V p-p L O PHASE V = ±5V S V = ±5V S –50 –90 V = +5V S –60 +1 –180 –70 0 –270 GAIN 2nd –80 –1 R = 150Ω L V = ±5V –90 –2 S 3rd –100 –3 R = 1kΩ L 2nd V = +5V S R = 1kΩ –110 L –4 –120 3rd –5 R = 150Ω L –6 1k 10k 100k 1M 10M 100M 1M 10M 100M 1G FREQUENCY – Hz FREQUENCY – Hz Figure 18. Closed-Loop Gain and Phase vs. Frequency, Figure 15. Harmonic Distortion vs. Frequency G = +1, R = 150 Ω L 1800 2000 V = ±5V 1800 1600 S R = 500Ω G = +10 L G = +10 1600 1400 1400 1200 G = –1 G = –1 1200 1000 G = +2 1000 G = +2 800 800 G = +1 600 G = +1 600 400 400 200 200 18 23 4567 1.5 2.5 3.5 4.5 5.5 6.5 7.5 SUPPLY VOLTAGE – ±Volts OUTPUT STEP SIZE – V p-p Figure 16. Slew Rate vs. Output Step Size Figure 19. Maximum Slew Rate vs. Supply Voltage 500mV 20ns 2V 20ns 100 100 V 90 V 90 IN IN V V OUT 10 10 OUT 0% 0% 500mV 2V Figure 20. Small Signal Pulse Response, Gain = +1, Figure 17. Large Signal Pulse Response, Gain = +1, (R = 2 kΩ, R = 150 Ω, V = ±5 V) (R = 2 kΩ, R = 150 Ω, V = ±5 V) F L S F L S REV. A –7– HARMONIC DISTORTION – dBc SLEW RATE – V/µs CLOSED-LOOP GAIN (NORMALIZED) – dB SLEW RATE – V/µs PHASE SHIFT – Degrees AD8013 50mV 20ns 2V 20ns 100 100 V V IN 90 90 IN V V 10 10 OUT OUT 0% 0% 500mV 2V Figure 21. Large Signal Pulse Response, Gain = +10, Figure 24. Large Signal Pulse Response, Gain = –1, R = 301 Ω, R = 150 Ω, V = ±5 V) (R = 698 Ω, R = 150 Ω, V = ±5 V) F L S F L S G = +10 G = –1 R = 150Ω 0 L 180 R = 150Ω L PHASE PHASE V = ±5V –90 90 S V = +5V V = ±5V S S V = +5V S +1 –180 +1 0 0 –270 0 –90 GAIN GAIN –1 –1 –2 V = +5V –2 S V = ±5V S –3 –3 V = ±5V S V = +5V S –4 –4 –5 –5 –6 –6 1M 10M 100M 1G 1M 10M 100M 1G FREQUENCY – Hz FREQUENCY – Hz Figure 22. Closed-Loop Gain and Phase vs. Frequency, Figure 25. Closed-Loop Gain and Phase vs. Frequency, G = +10, R = 150 Ω G = –1, R = 150 Ω L L 50mV 20ns 500mV 20ns 100 100 V 90 V 90 IN IN V 10 V 10 OUT OUT 0% 0% 500mV 500mV Figure 26. Small Signal Pulse Response, Gain = –1, Figure 23. Small Signal Pulse Response, Gain = +10, (R = 301 Ω, R = 150 Ω, V = ±5 V) (R = 698 Ω, R = 150 Ω, V = ±5 V) F L S F L S –8– REV. A CLOSED-LOOP GAIN (NORMALIZED) – dB PHASE SHIFT – Degrees CLOSED-LOOP GAIN (NORMALIZED) – dB PHASE SHIFT – Degrees AD8013 To estimate the –3 dB bandwidth for closed-loop gains of 2 or greater, for feedback resistors not listed in the following table, G = –10 the following single pole model for the AD8013 may be used: R = 150Ω 180 L PHASE G V = ±5V S 90 ACL . 1+ SC (R + Gn rin) V = +5V S T F +1 0 where: C = transcapacitance > 1 pF T 0 –90 R = feedback resistor GAIN F –1 G = ideal closed loop gain –2  R  F 1+ Gn =   = noise gain –3 V = ±5V R   S G V = +5V S rin = inverting input resistance > 150 Ω –4 ACL = closed loop gain –5 The –3 dB bandwidth is determined from this model as: –6 1M 10M 100M 1G 1 FREQUENCY – Hz f . 3 2 π C (R + Gn rin) T F Figure 27. Closed-Loop Gain and Phase vs. Frequency, This model will predict –3 dB bandwidth to within about 10% G = –10, R = 150 Ω L to 15% of the correct value when the load is 150 Ω and V = S ±5 V. For lower supply voltages there will be a slight decrease in General bandwidth. The model is not accurate enough to predict either The AD8013 is a wide bandwidth, triple video amplifier that the phase behavior or the frequency response peaking of the offers a high level of performance on less than 4.0 mA per AD8013. amplifier of quiescent supply current. The AD8013 uses a It should be noted that the bandwidth is affected by attenuation proprietary enhancement of a conventional current feedback due to the finite input resistance. Also, the open-loop output architecture, and achieves bandwidth in excess of 200 MHz with resistance of about 12 Ω reduces the bandwidth somewhat when low differential gain and phase errors, making it an extremely driving load resistors less than about 250 Ω. (Bandwidths will efficient video amplifier. be about 10% greater for load resistances above a few hundred The AD8013’s wide phase margin coupled with a high output ohms.) short circuit current make it an excellent choice when driving Table I. –3 dB Bandwidth vs. Closed-Loop Gain and Feedback any capacitive load. High open-loop gain and low inverting Resistor, R = 150 Ω (SOIC) input bias current enable it to be used with large values of L feedback resistor with very low closed-loop gain errors. V – Volts Gain R – Ohms BW – MHz S F It is designed to offer outstanding functionality and performance ±5 +1 2000 230 at closed-loop inverting or noninverting gains of one or greater. +2 845 (931) 150 (135) +10 301 80 Choice of Feedback & Gain Resistors –1 698 (825) 140 (130) Because it is a current feedback amplifier, the closed-loop band- –10 499 85 width of the AD8013 may be customized using different values +5 +1 2000 180 of the feedback resistor. Table I shows typical bandwidths at +2 887 (931) 120 (130) different supply voltages for some useful closed-loop gains when +10 301 75 driving a load of 150 Ω. –1 698 (825) 130 (120) The choice of feedback resistor is not critical unless it is –10 499 80 important to maintain the widest, flattest frequency response. Driving Capacitive Loads The resistors recommended in the table are those (chip When used in combination with the appropriate feedback resistors) that will result in the widest 0.1 dB bandwidth without resistor, the AD8013 will drive any load capacitance without peaking. In applications requiring the best control of bandwidth, oscillation. The general rule for current feedback amplifiers is 1% resistors are adequate. Package parasitics vary between the that the higher the load capacitance, the higher the feedback 14-pin plastic DIP and the 14-pin plastic SOIC, and may result resistor required for stable operation. Due to the high open-loop in a slight difference in the value of the feedback resistor used to transresistance and low inverting input current of the AD8013, achieve the optimum dynamic performance. Resistor values and the use of a large feedback resistor does not result in large closed- widest bandwidth figures are shown in parenthesis for the SOIC loop gain errors. Additionally, its high output short circuit current where they differ from those of the DIP. Wider bandwidths than makes possible rapid voltage slewing on large load capacitors. those in the table can be attained by reducing the magnitude of For the best combination of wide bandwidth and clean pulse the feedback resistor (at the expense of increased peaking), response, a small output series resistor is also recommended. while peaking can be reduced by increasing the magnitude of Table II contains values of feedback and series resistors which the feedback resistor. result in the best pulse responses. Figure 29 shows the AD8013 Increasing the feedback resistor is especially useful when driving driving a 300 pF capacitor through a large voltage step with large capacitive loads as it will increase the phase margin of the virtually no overshoot. (In this case, the large and small signal closed-loop circuit. (Refer to the section on driving capacitive pulse responses are quite similar in appearance.) loads for more information.) REV. A –9– CLOSED-LOOP GAIN (NORMALIZED) – dB PHASE SHIFT – Degrees AD8013 R F As noted in the warning under “Maximum Power Dissipation,” 1.0µF +V S a high level of input overdrive in a high noninverting gain circuit 0.1µF can result in a large current flow in the input stage. Though this R G current is internally limited to about 30 mA, its effect on the 4 15Ω total power dissipation may be significant. AD8013 V O 1.0µF R S C 11 V L IN High Performance Video Line Driver 0.1µF R T At a gain of +2, the AD8013 makes an excellent driver for a –V S back terminated 75 Ω video line (Figures 31, 32, and 33). Low differential gain and phase errors and wide 0.1 dB bandwidth Figure 28. Circuit for Driving a Capacitive Load can be realized. The low gain and group delay matching errors ensure excellent performance in RGB systems. Figures 34 and Table II. Recommended Feedback and Series Resistors vs. 35 show the worst case matching. Capacitive Load and Gain R R F G R – Ohms S +V S C – pF R – Ohms G = 2 G ≥ 3 L F 0.1µF 75Ω 20 2k 25 15 4 CABLE 75Ω 75Ω 50 2k 25 15 AD8013 V CABLE OUT 100 3k 20 15 11 0.1µF 75Ω V IN 200 4k 15 15 75Ω 300 6k 15 15 –V S ≥500 7k 15 15 Figure 31. A Video Line Driver Operating at a Gain of +2 (R = R from Table I) 500mV F G 50ns 100 90 V IN G = +2 R = 150Ω 0 L PHASE V = ±5V –90 S V = +5V S +1 –180 10 0 –270 V OUT GAIN 0% –1 V = ±5V 1V S –2 –3 V = +5V Figure 29. Pulse Response Driving a Large Load Capacitor. S –4 C = 300 pF, G = +2, R = 6k, R = 15 Ω L F S –5 Overload Recovery –6 The three important overload conditions are: input common- 1M 10M 100M 1G FREQUENCY – Hz mode voltage overdrive, output voltage overdrive, and input current overdrive. When configured for a low closed-loop gain, Figure 32. Closed-Loop Gain & Phase vs. Frequency the amplifier will quickly recover from an input common- for the Line Driver mode voltage overdrive; typically in under 25 ns. When con- figured for a higher gain, and overloaded at the output, the G = +2 recovery time will also be short. For example, in a gain of +10, R = 150Ω L with 15% overdrive, the recovery time of the AD8013 is about +0.2 20 ns (see Figure 30). For higher overdrive, the response is +0.1 somewhat slower. For 6 dB overdrive, (in a gain of +10), the recovery time is about 65 ns. 0 –0.1 V = ±5V S 500mV 50ns –0.2 V = +5V 100 S –0.3 V 90 IN –0.4 –0.5 1M 10M 100M 1G 10 V FREQUENCY – Hz OUT 0% 5V Figure 33. Fine-Scale Gain Flatness vs. Frequency, G = +2, R = 150 Ω L Figure 30. 15% Overload Recovery, G = +10 (R = 300 Ω, F R = 1 kΩ, V = ±5 V) L S –10– REV. A CLOSED-LOOP GAIN (NORMALIZED) – dB NORMALIZED GAIN – dB PHASE SHIFT – Degrees AD8013 1.5 V +5V I 8k G = +2 1.0 R = 150Ω L TO DISABLE PIN 0.5 4k 10k 0 –5V V = +5V S V HIGH => AMPLIFIER ENABLED I –0.5 V LOW => AMPLIFIER DISABLED I V = ±5V S –1.0 Figure 36. Level Shifting to Drive Disable Pins on Dual Supplies –1.5 The AD8013’s input stages include protection from the large –2.0 differential input voltages that may be applied when disabled. 1M 10M 100M 1G Internal clamps limit this voltage to about ±3 V. The high input to FREQUENCY – Hz output isolation will be maintained for voltages below this limit. Figure 34. Closed-Loop Gain Matching vs. Frequency 3:1 Video Multiplexer 10 Wiring the amplifier outputs together will form a 3:1 mux with G = +2 excellent switching behavior. Figure 37 shows a recommended 8 R = 150Ω L configuration which results in –0.1 dB bandwidth of 35 MHz 6 and OFF channel isolation of 60 dB at 10 MHz on ±5 V V = +5V S 4 supplies. The time to switch between channels is about 50 ns. 2 Switching time is virtually unaffected by signal level. V = ±5V S DELAY 665Ω 845Ω 1.0 G = +2 +V S 0.5 R = 150Ω L V = ±5V S 0 6 4 84Ω V = +5V S 7 –0.5 DELAY MATCHING 1 5 V 1 IN –1.0 75Ω 100k 1M 10M 100M FREQUENCY – Hz DISABLE 1 665Ω 845Ω Figure 35. Group Delay and Group Delay Matching vs. Frequency, G = +2, R = 150 Ω 75Ω L 13 CABLE 84Ω 14 V OUT 12 2 Disable Mode Operation V 2 75Ω IN 75Ω Pulling the voltage on any one of the Disable pins about 1.6 V up from the negative supply will put the corresponding DISABLE 2 amplifier into a disabled, powered down, state. In this 665Ω 845Ω condition, the amplifier’s quiescent current drops to about 0.3 mA, its output becomes a high impedance, and there is 9 84Ω a high level of isolation from input to output. In the case of 8 11 the gain of two line driver for example, the impedance at the 10 V 3 3 IN 75Ω output node will be about the same as for a 1.6 kΩ resistor (the feedback plus gain resistors) in parallel with a 12 pF –V S DISABLE 3 capacitor and the input to output isolation will be about 66 dB at 5 MHz. Figure 37. A Fast Switching 3:1 Video Mux (Supply Leaving the Disable pin disconnected (floating) will leave Bypassing Not Shown) the corresponding amplifier operational, in the enabled state. The input impedance of the disable pin is about 40 kΩ 200ns 500mV in parallel with a few picofarads. When driven to 0 V, with 100 the negative supply at –5 V, about 100 μA flows into the 90 disable pin. When the disable pins are driven by complementary output CMOS logic, on a single 5 V supply, the disable and enable times are about 50 ns. When operated on dual supplies, 10 level shifting will be required from standard logic outputs to 0% the Disable pins. Figure 36 shows one possible method 5V which results in a negligible increase in switching time. Figure 38. Channel Switching Characteristic for the 3:1 Mux REV. A –11– GAIN MATCHING – dB GROUP DELAY – ns AD8013 2:1 Video Multiplexer The AD8013 can be used to build a circuit for switching between Configuring two amplifiers as unity gain followers and using the any two arbitrary gains while maintaining a constant input third to set the gain results in a high performance 2:1 mux impedance. The example of Figure 41 shows a circuit for switching (Figures 39 and 40). This circuit takes advantage of the very low between a noninverting gain of 1 and an inverting gain of 1. The crosstalk between Channels 2 and 3 to achieve the OFF channel total time for channel switching and output voltage settling is isolation shown in Figure 40. This circuit can achieve about 80 ns. differential gain and phase of 0.03% and 0.07° respectively. 698Ω 698Ω +5V R1 6 2k 13 2kΩ 4 14 5 7 2 1 13 12 14 2 R3 V A IN 100Ω 10Ω DIS 1 V 15Ω IN 12 DISABLE 5 V 1k OUT DIS 3 50Ω R4 7 1 V OUT 10 V B 10Ω 10 IN 3 6 8 3 R5 845Ω 9 8 11 9 R2 1k 3 2kΩ R6 –5V 845Ω DISABLE 845Ω 845Ω Figure 39. 2:1 Mux with High Isolation and Low Differential Gain and Phase Errors Figure 41. Circuit to Switch Between Gains of –1 and +1 2 1 200ns 500mV 500mV GAIN 0 100 90 –1 –2 –30 –3 –40 –4 10 0% –50 –5 FEEDTHROUGH 5V –60 –6 –70 –7 –8 –80 Figure 42. Switching Characteristic for Circuit of Figure 41 1M 10M 100M 1G FREQUENCY – Hz Figure 40. 2:1 Mux ON Channel Gain and Mux OFF Channel Feedthrough vs. Frequency Gain Switching OUTLINE DIMENSIONS Dimensions shown in inches and (mm). 14-Lead SOIC (R-14) 14-Lead Plastic DIP (N-14) 0.3444 (8.75) 0.795 (20.19) 0.3367 (8.55) 0.725 (18.42) 14 8 14 8 0.280 (7.11) 0.1574 (4.00) 0.2440 (6.20) 0.240 (6.10) 1 7 17 0.1497 (3.80) 0.2284 (5.80) 0.325 (8.25) 0.195 (4.95) 0.060 (1.52) 0.300 (7.62) PIN 1 0.115 (2.93) 0.015 (0.38) PIN 1 0.0688 (1.75) 0.0196 (0.50) x 45° 0.0098 (0.25) 0.0532 (1.35) 0.210 (5.33) 0.0099 (0.25) MAX 0.0040 (0.10) 0.130 (3.30) 0.160 (4.06) MIN 0.115 (2.93) 8° 0.015 (0.381) 0.0500 0.0192 (0.49) 0° SEATING SEATING 0.0500 (1.27) 0.022 (0.558) 0.100 0.070 (1.77) 0.0098 (0.25) 0.008 (0.204) (1.27) 0.0138 (0.35) PLANE PLANE (2.54) BSC 0.0160 (0.41) 0.014 (0.356) 0.045 (1.15) 0.0075 (0.19) BSC –12– REV. A CLOSED-LOOP GAIN – dB FEEDTHROUGH – dB PRINTED IN U.S.A. C2084–18–10/95